ZERON -A Super Simple QRP Dual Band Multimode Transceiver. -1

I always admire the versatility of digital chips and use them in many analog designs as well. My curiosity for their implementation in analog circuits threads back to mid nineties, following an article in QEX magazine. Later I used them in many designs and their inexpensiveness and outstanding performance impressed me to the added advantage of compact designs. this resulted in many successful designs including "DIGIRIG" and "NANO" transceivers. After publication of "DIGIRIG" (SPRAT 84 Page 10), late Bill Currie VK3AWC wrote me mentioning that he actually liked this kind of H.F. designs and  he was doing similar sort of experiments.Later, between 1992 and 1999 we worked on many similar designs including receivers, multi-band mixer type VFOs and even weaver's exciter. This  inexpensive diminutive multi-mode, dual band (80/40M) rig is designed  on the similar lines, and is aimed  for the novice amateurs. The schematic diagram shows the transmitter side of the transceiver.

As I already mentioned in one of my previous post that QRP is synonymous with simplicity and minimalism, the present design illustrates it very well. The audio from the electret mic is amplified by a simple single stage amplifier wired around BC548B. This is coupled to a differential AF amplifier wired around a pair of LM 386N, via an audio transformer having 1K:1K+1K impedance taps. This sort of coupling transformer isn't  difficult to find and even digikey  stocks it.

A 74HC4053 functions as a mixer. I used it as it is very cheap, easier to find and requires fewer peripheral components. S1, a double pole three way rotary switch acts as a mode switch that chooses the desired mode of operation. The output of the mixer is amplified by a single IRF530 and is fed to the antenna through an LPF. Too simple to say anything about. Do you still feel the need to burn your bucks to get yourself on the air! In my next post I will discuss the RX and VFO units of this ultra simple project.

PIXET- AN H.F. CW/SSB TRANSMITTER-III

More time is spent on designing the RF power amplifier module than any other part of "PIXET" project. I started with a design based on single ended topology, but later gave up after cooking several IRF510s. Then I moved on to the present design topology for its inherent immunity to harmonics with added benefits of lesser heat and added stability.

Seperate biasing networks for both IRF510s are used. All tranformers are wound over common T50-43 toroids except the PA output transformer. This is wound with three turns each on BN61-202 twin hole large balun core. Mind phasing for the primary winding only. Winding is with 24 SWG copper enamelled wire. 

L2 is wound on a small pignose balun core having 10 turns of 30 SWG. I made it as a choke and a fuse as well but if you wish to use an extra fuse than you can do so. 

Though HEXFETS of IRF series are commonly used in HF power amplifiers but their gate capacitance is always to large and shunts down a considerable drive, especially on 20 meter and above. Thus the performance of HEXFET power amplifiers is usually, noticeably poorer on the 20 meter and above. L3 and L4 serves the purpose to tune this capacitance on desired band of operation. I used about seventeen turns of 26 SWG wound over a lead pencil and then removed as a self standing air core coil. You can either spread or compress the turns to make refinement in performance, or may add or remove turns as required on the band of operation. This just serves a starting point. These coils tunes the gate capacitance on the operting frequency of the band and the effect of input capacitance is thus nulls out, making the amplifier perform better on higher frequency bands.

PIXET- AN H.F. CW/SSB TRANSMITTER-II

In my last post I presented the circuit of the exciter unit of the "PIXET" transmitter, a companion unit for "PIXER" receiver described earlier. In this second part of it I am describing the RF speech processor unit, the second mixer and pre driver circuitry.

Basics of RF speech processing: Imagine a power amplifier designed for 10 W pep driven by a mean ssb signal, which at least will be down by 6 db below the peak. This means a minimum output power of 2.5 watts and a resulting S-meter reading of one step down from the peak. Equalizing the dynamic range of the modulating signal will result in a better effectivity of the power amplifier as this will raise the "mean" output power. Even if this might not be directly visible at the receiver S meter, the compression of the dynamic range will increase the readability and the SNR at the receiving side. In practice, it could be proven that a – moderate - clipping limit of 20 db virtually simulates an 80 watts transmitter while, in reality, the pep output is only 10 watts. Let's understand the root of this philosophy.

The Felcher-Munson Philosophy:  Grokking this theory is a bit beyond my brain right now, but the Fletcher–Munson curves are one of many sets of equal-loudness contours for the human ear, determined experimentally by Harvey Fletcher and Wilden A. Munson, and reported in a 1933 paper entitled "Loudness, its definition, measurement and calculation". The first research on the topic of how the ear hears different frequencies at different levels was conducted by Fletcher and Munson in 1933. In 1937 they created the first equal-loudness curves. Until recently, it was common to see the term 'Fletcher–Munson' used to refer to equal-loudness contours generally, even though a re-determination was carried out by Robinson and Dadson in 1956, which became the basis for an ISO 226 standard.

                              

It is now better to use the generic term "equal-loudness contours", especially as a recent survey by ISO redefined the curves in a new standard. According to the ISO report, the Robinson–Dadson results were the odd one out, differing more from the current standard than did the Fletcher Munson curves. The report states that it is fortunate that the 40-phon Fletcher–Munson curve on which the A-weighting standard was based turns out to have been in agreement with modern determinations. The article also comments on the large differences apparent in the low-frequency region, which remain unexplained. Possible explanations are:

1. The equipment used was not properly calibrated.

2. The criteria used for judging equal loudness at different frequencies had differed.

3. Subjects were not properly rested for days in advance, or were exposed to loud noise in traveling to the tests which tensed the tensor tympani and stapedius muscles controlling low-frequency mechanical coupling.

Thus equal-loudness curves derived using headphones are valid only for the special case of what is called side-presentation, which is not how we normally hear. Real-life sounds arrive as planar wavefronts, if from a reasonably distant source. If the source of sound is directly in front of the listener, then both ears receive equal intensity, but at frequencies above about 1 kHz the sound that enters the ear canal is partially reduced by the masking effect of the head, and also highly dependent on reflection off the pinna (outer ear). Off-centre sounds result in increased head masking at one ear, and subtle changes in the effect of the pinna, especially at the other ear. This combined effect of head-masking and pinna reflection is quantified in a set of curves in three-dimensional space referred to as head-related transfer functions (HRTFs). Frontal presentation is now regarded as preferable when deriving equal-loudness contours, and the latest ISO standard is specifically based on frontal and central presentation.

The A-weighting curve—in widespread use for noise measurement—is said to have been based on the 40-phon Fletcher–Munson curve. However, research in the 1960s demonstrated that determinations of equal-loudness made using pure tones are not directly relevant to our perception of noise. This is because the cochlea in our inner ear analyzes sounds in terms of spectral content, each "hair-cell" responding to a narrow band of frequencies known as a critical band. The high-frequency bands are wider in absolute terms than the low frequency bands, and therefore "collect" proportionately more power from a noise source. However, when more than one critical band is stimulated, the outputs of the brain sum the various bands to produce an impression of loudness. For these reasons Equal-loudness curves derived using noise bands show an upwards tilt above 1 kHz and a downward tilt below 1 kHz when compared to the curves derived using pure tones.

BBC Research conducted listening trials in an attempt to find the best weighting curve and rectifier combination for use when measuring noise in broadcast equipment, examining the various new weighting curves in the context of noise rather than tones, confirming that they were much more valid than A-weighting when attempting to measure the subjective loudness of noise. This work also investigated the response of human hearing to tone-bursts, clicks, pink noise and a variety of other sounds that, because of their brief impulsive nature, do not give the ear and brain sufficient time to respond. 

What does that actually mean: The way to read this graph is as follows: look at the blue curve at the 1 kHz / 40 dB point. Now follow the curve towards the left until you reach 50 Hz on the horizontal axis. You should now read about 70 dB on the vertical axis. In essence, this states that in order for a 50 Hz tone to be perceived as loud as a 1 kHz tone is at 40 dB, it needs to be played at 70 dB. That’s 30 dB difference! A similar thing happens when you move into the high frequencies. A 10 kHz tone needs to be played at about 55 dB to be perceived at the same loudness level. Notice that this difference in loudness evens out as the volume increases (the curves higher up in the figure), for example at 100 dB, the curves have flatten out considerably, meaning the perceived loudness difference between tones at different frequencies decreases. There are two important things to take away from these curves:

1. We are less sensitive to low and high frequencies, we hear mid frequencies more prominently (especially between 1-5 kHz)

2. As the volume increases, this perceived loudness difference between the frequencies diminishes.

However, this made the basis of one of the pioneering developments in low power DX voice communication in which the high amplitude vocals are compressed for an even distribution of power over the usable bandwidth. Based upon this research; in HF-SSB radio technology in the era of late sixties, became a dependable method of modifying the speech waveform in the transmitter to produce a marked improvement in the signal-to-noise ratio at the receiver without also causing any significant increase in distortion products, either in-band or out-of-band. Since RF speech processing was the key to the performance of low-power HF-SSB radio sets - and is now recognized almost as a sine qua non in SSB transmitters - the principles involved will be described briefly. Typically, unprocessed speech has a ratio of instantaneous peak to average power of about 16dB (see Pictures below:

Unprocessed Signal

Processed Signal

In a peak-power-limited system, such as an SSB transmitter, this represents a considerable loss of potential output power, so some means of compressing the dynamic range of the speech signal is required before transmission. It is now well known that clipping (or hard limiting) the peaks of an SSB waveform, and then filtering by a second bandpass filter similar to that in a filter-type SSB generator to remove the resulting harmonic and high-order products, can markedly improve the articulation index of the transmitted signal. Methods of doing this were just being developed around mid sixties. Though modern day transmitters implement compression through DSP techniques using digital algorithms but this project describes a very elegant alternative compressor design like one of the yore.

PIXET Speech processor: The following schematic illustrates the complete circuit diagram of the speech processor and the second X-tal filter:

The SSB signal from the exciter unit is compressed using   diode D1 and the base collector junction of transistor Q1. While developing the clipper circuit I initially employed two back to back diodes. But at such a low level signals adequate level of clipping demands for special hot carrier diodes like HP 5082-2811; which are both expensive and hard to find for an average experimenter. Consequently I zeroed my choice for this simple and effective alternative circuit. Both threshold and gain controls are required to be adjusted carefully. An oscilloscope is quite invaluable tool to do this precious adjustment but in case of its non- availability on the air adjustment also provides convincing results. You can feed the output signal of the transmitter into a dummy load and the adjustments of required gain and proper compression can be done by hearing the signal in a nearby receiver.

Second mixer and pre-driver: There is nothing special to explain in this section. An SBL-1 mixer is used but a home brew variety of double balanced diode mixer will perform equally well. Attempts are made to terminate all mixer ports towel defined 50 ohms impedance to ensure optimum IMD performance. The band pass filter is W7ZOI design which can be scaled to other bands of interest if desired. I tested the transmitter on 14MHz band by feeding VFO signal from 18.43 to 18.78 Mhz. The higher side VFO injection automatically puts you on the right side of sideband i.e. USB. For lower sideband operation as on 80 and 40 meters, lower side VFO injection could be used. Or you can just put up a two banded with single IF just by band switching VFO signal. In that case sideband selection would be automatic for both bands, but the VFO tuning will be in opposite direction. This will help a newbie to assemble a two bander with minimum effort and cost.

In the next post I will describe the R.F. linear  amplifier for the "PIXET" transmitter.

PIXET- AN H.F. CW/SSB TRANSMITTER-I

It all began with the nonavailability of expensive mixer ICs. I homebrewed my own mixer IC (parts shown in dotted square), and as I had already described it in my blog post of 9th February. Keen  to having made the "PIXER" superhet receiver and other projects based on it...........finally I started to build a DSB rig around it but over the last couple of weeks, design evolved with a bit of tinkring here and there and ultimately it took shape of a very nice companian H.F. CW/SSB transmitter to "PIXER"; having a clean spectral purity and a a side band suppression of around 58 dB. Indeed great for a thing developed around common off the shelf components and right on your kitchen table......and all that with a home brewed X-tal filter!!! The schematic of the exciter unit is as under:

The circuit of PIXET exciter is quite simple in itself and requires no description. I have used components that were available on hand. Especially the transistors. You can substitute them with any general purpose medium power ones having adequate gain bandwidth product. Though matched pair of transistors has been used in the balanced mixer but R6 has been included for refining the circuit balance. It can be carefully adjusted for minimum carrier during no audio signal present at the microphone input. A simple diode gate provides the CW keying. Could there be anything simpler than that....? And that's the real essence of QRP home brewing.....!!!

PIXER- An empirical H.F. Superhet Receiver.-II

In the last post I described the IF and product detector of an empirical superhet receiver developed around home made ICs as described in my post of 9th February (parts shown in dotted squares). In this post I am going to describe the front end design of this receiver.

Initially, I decided to wire the front end using another home brewed mixer IC. But out of an amateur's true spirit I decided to do experiment a bit more. After a receiver's sensitivity, the next requirement is its ability to discern weak signals in the presence of strong signals in its pass-band. This is known as dynamic range of the receiver.

There are several types of dynamic range. The first one, and probably the easiest to understand-"AGC range"-concerns whether a receiver is capable of maintaining a constant audio output level over a large input-signal amplitude range. The traditional school of thought requires AGC action to commence at about 3µV, leading to a condition where signals that produce an excellent signal-to-noise ratio may show absolutely no S-meter indication-a most undesirable effect. The reason for this is inappropriate receiver gain distribution-generally, a lack of gain at the IF. Maintaining constant audio output must involve gain control at the receiver's IF, and possibly even at its input.

IMD Dynamic Range:  The output of a linear stage tracks the input signal decibel by decibel, with every 1-dB change in its input signal corresponding to an identical 1-dB output change. This is the stage's first-order response. Because no device is perfectly linear, however, two or more signals applied to it intermodulate to some degree, generating sum and difference frequencies. These intermodulation distortion (IMD) products occur at frequencies and amplitudes that depend on the order of the IMD response as follows:

•Second-order IMD products change 2 dB for every decibel of input-signal change, and appear at frequencies that result from the simple addition and subtraction of input-signal frequencies. For example, assuming that its input bandwidth is sufficient to pass them, an amplifier subjected to signals at 6 and 8 MHz will produce second-order IMD products at 2 MHz (8 - 6) and 14 MHz (8 + 6).

•Third-order IMD products change 3 dB for every decibel of input-signal change, and appear at frequencies corresponding to the sums and differences of twice one signal's frequency plus or minus the frequency of another. Assuming that its input bandwidth is sufficient to pass them, an amplifier subjected to signals at 14.02 MHz (f1) and 14.04 MHz (f2) produces third-order IMD products at 14.00 (2f1 - f2), 14.06 (2f2 - f1), 42.08 (2f1 + f2) and 42.10 (2f2 + f1) MHz. The subtractive products (the 14.00 and 14.06-MHz products in this example) are close to the desired signal and can cause significant interference. This is why our receivers' third-order IMD performance is so important. It can be seen that the IMD order determines how rapidly IMD products change level per unit change of input level. Nth-order IMD products therefore change by n dB for every decibel of input-level change. IMD products at orders higher than three can and do occur in communication systems, but the second- and third-order products are most important in receiver front ends.

Intercept Point: The second type of dynamic range concerns the receiver's intercept point, sometimes simply referred to as input intercept. Intercept point is typically measured by applying two or three closely spaced signals to the antenna input, tuning the receiver to count the number of resulting spurious responses, and measuring their level relative to the input signal.

Because a device's IMD products increase more rapidly than its desired output as the input level rises, it might seem that steadily increasing the level of multiple signals applied to an amplifier would eventually result in equal desired-signal and IMD levels at the amplifier output. Real devices are incapable of doing this, however. At some point, every device overloads, and changes in its output level no longer equally track changes at its input. The device is then said to be operating in compression. Pushing the process to its limit ultimately leads to saturation, at which point input-signal increases no longer increase the output level.

The power level at which a device's second-order IMD products equal its first-order output (a point that must be extrapolated because the device is in compression by this point) is its second-order intercept point. Likewise, its third-order intercept point is the power level at which third-order responses equal the desired signal. The following figure represents these relationships:

A linear stage's output tracks its input decibel by decibel on a 1:1 slope-its first-order response. Second-order intermodulation distortion (IMD) products produced by two equal-level input signals ("tones") rise on a 2:1 slope-2 dB for every 1 dB of input increase. Third-order IMD products likewise increase 3 dB for every 1 dB of increase in two equal tones. For each IMD order n, there is a corresponding intercept point IPn at which the stage's first-order and nth order products are equal in amplitude. The first order output of real amplifiers and mixers falls off (the device overloads and goes into compression) before IMD products can intercept it, but intercept point is nonetheless a useful, valid concept for comparing radio system performance. The higher an amplifier or mixer's intercept point, the stronger the input signals it can handle without overloading. The input and output powers shown are for purposes of example; every receiver exhibits its own particular IMD profile.

Input filtering can improve second-order intercept point; device non-linearities determine the third, fifth and higher-odd number intercept points. In pre-amplifiers, third-order intercept point is directly related to dc input power; in mixers, to the local-oscillator power applied.

Intercept point can be confusing because it can be specified in terms of input or output power. Intercept point should be referred to device output because that's where the trouble occurs, but input intercept is commonly given. Therefore, if an amplifier or a mixer has a particular intercept point-let's say +30 dBm at 10 dB gain-and then its gain is increased by an additional 10 dB, its dynamic range decreases by the amount of the gain.

Thus the first requirement for a receiver's front-end to have a good dynamic range, is a good mixer. My choice thus zeroed on the simple diode ring mixer which already has gained popularity among amateur fraternity. Double balanced mixers are a form of what is termed a "reversing switch mixer." Reversing switch mixers operate by using electronic switches in a bridge formation to reverse the input RF signal under the action of the local oscillator used as a square wave switching signal. They normally offer significant advantages over analogue mixers for radio communications and general RF design applications as they are able to offer better levels of dynamic range and noise. In view of this fact, they are normally used in high performance applications where noise and dynamic range are of importance - e.g. in the front end of a radio receiver or spectrum analyzer.

Although there are comparatively few components in a double balanced mixer, their individual performance is crucial to the performance of the RF mixer as a whole. Normally Schottky barrier diodes are used for the diode ring. They offer a low on resistance and they also have a good high frequency response. Ordinary signal diodes may be used for low performance applications, although the cost difference is small. It is found that the diode forward voltage drop for the diodes determines the optimum local oscillator drive level. RF mixers requiring to handle a high RF input level will need a correspondingly high LO input level. As a rule of thumb the LO signal level should be a minimum of 20dB higher than either the RF or IF signals. This ensures that the LO signal rather than the RF or IF signals switch the RF mixer, and this is a key element in reducing intermodulation distortion, IMD, and also maximising the dynamic range.

To increase the required drive level, it is possible to place multiple diodes in each leg. The most common LO drive level for a double balanced mixer is probably +7dBm. However they can be obtained with a variety of drive levels. Values of 0, +3, +7, +10, +13, +17, +23, and +27 dBm are normally used.

I decided to use a home brew mixer using common inexpensive diodes in the front-end, the schematic is as under:

The input signals is passed through a double tuned band pass circuit made around slug tuned, self wound inductors. I included a low pass circuit owing to ring mixers ability to respond to strong, harmonic signals. The given values are chosen for forty meter band, but they can easily be scaled for other frequency bands, too. Or even a multi band operation is also possible using suitable band switching. The home made mixer uses four matched diodes and two RF transformers wound on pig-nose balun cores. Transformers have thirteen trifilars turns. I used inexpensive diodes in the mixer but they behaved extremly well. Any 2.5MHz VFO capable of delivering reasonable power can be used. Amateur literature is already full of several circuits. For a good input intercept to be maintained it is important to properly terminate all mixer ports. IF port of the mixer is terminated in a post mixer amplifier that terminates the mixer output to an appropriate impedance, required to maintain a good IP3. The IF signal is then routed to a crystal filter through a post mixer amplifier, as shown below:

The post mixer amplifier uses noiseless inductive feedback. I used 2N3866 as it was available, but 2N4427, BFW16A etc. will seem to work equally well. Keep the transistor leads as short as possible and try to use ferrite beads in the collector lead to avoid spurs. I used a variable bandwidth X-Tal filter whose bandwidth can be controlled with R12. Cheap color burst X-Tals of 4.43 MHz are used for the x-tal filter. On twenty meters and above a low noise amplifier ahead of mixer is recommended to achive optimal noise figure.

The overall performance is amazingly good and despite designed around common off the shelf type inexpensive components it performs really well, far better than many commercial receivers.

PIXER- An empirical H.F. Superhet Receiver.-I

In my last posts I discussed an IF amplifier and a simple direct conversion receiver. Why not try building an empirical H.F. Station Superhet using these building blocks? It can make an excellent basis for both fun and learning.

Receiver basics:  It is imperative that certain criteria must be met in the design of even a basic receiver even of a simplest form. These include sufficient gain to provide reasonable sensitivity and selectivity.

The first requirement is to provide reasonable gain. The signal levels from the antenna are quite low, while reasonable power to drive a speaker is usually required. If a signal of 0.1uV is assumed to be received by a receiver with 50 ohm antenna impedance, the power available to the receiver will be 2 X 10^-16 watts. If we would have to produce a discernible signal at the output of the receiver by driving an 8 ohm speaker at 10 X 10^-4 watts or say one milliwatt , a gain of  (10 X 10^-4 )/( 2 X 10^-16 ) is thus desirable. This translates to 5 X 10^-12 or nearly 127 dB of the gain required. This is a typical figure for many receivers. Since signals having lesser power can be copied easily by sensitive speakers or headphones and/or full volume reception is usually not desirable, a little less gain is thus suffice our needs. The absolute minimum figure thus comes down to around 100 dB, you can say.

The second requirement is to process the input signal to produce a tangible output signal. This process is called demodulation or detection. This process differs for the signals carrying information in different forms called radio modes. And the third requirement of any receiver is to separate the desired signal from the crowd of un-wanted signals. This quality is referred as selectivity of the receiver which will be achieved using a home made x-tal filter. Shown below is the block diagram of a common superhet receiver:

The superhet receiver can really be seen in two parts: From IF amplifier onwards, it is a fixed frequency receiver, a receiver pre-tuned and optimized for the reception of a signal on the IF frequency. The RF amplifier and mixer/oscillator receive signals from the antenna and then convert them to the frequency of this optimum receiver - to the IF frequency. It is in the RF amplifier and mixer/oscillator sections of the receiver where the actual operator adjustment and tuning for the selection or choice of received signal takes place.

The choice of Intermediate Frequency:  There are two conflicts with the choice of the IF Frequency. A low intermediate frequency brings the advantage of higher stage gain and higher selectivity using high-Q tuned circuits. Sharp pass-bands are possible for narrow-band working for CW and SSB reception. A high intermediate frequency brings the advantage of a lower image response. The image frequency problem can be seen in this example.

Consider a receiver for 10 MHz using an IF frequency of 100 kHz. The local oscillator will be on either 10.1 MHz - i.e. 100 kHz higher than the required input signal - or on 9.9 MHz. We will consider the 10.1 MHz case - but the principles are the same for the case where the oscillator is LOWER in frequency than the wanted signal frequency. . Because of the way that mixers work, a signal at 10.2 MHz will also be received. This is known as the "image" frequency.

The image rejection of a superhet receiver can be improved by having more tuned circuits set to the required input frequency, such as more tuned circuits in the RF amplifier ahead of the mixer. This brings practical construction difficulties. Another solution is to choose a high IF frequency so that the required received frequency and the image frequency are well separated in frequency.

Choosing an IF of 2 MHz for the 10 MHz receiver would put the local oscillator at 12 MHz, the image frequency then being at 14 MHz. When receiving a signal at 10 MHz, it is easier to reject a signal at 14 MHz (the image in the 2 MHz IF case) than at 10.2 MHz (the image in the 100kHz IF case).

(Note that the Image Frequency is TWICE the IF Frequency removed from the WANTED signal frequency - on the same side of the wanted frequency as the oscillator).

PIXER H.F. Superhet Receiver:  The design of our empirical receiver is largely based upon the IF strip using home brewed mixer ICs, from my old posts of 9th and 24th March,2016. This IF strip is designed around a handful of "off the shelf" category inexpensive components and represents a quite respectable gain of about 68db on 4.43 MHz along with good AGC characteristics. I have chosen this frequency for my IF since color burst  X-tals for this frequency are easily available and are quite cheap. The complete circuit of IF amplifier is given below:

A product detector has to follow the IF amplifier. I decided to use the simple DC receiver described in my last post as the product detector. Minor modifications were needed and this work was not difficult as all the construction was done, using dead bug technique. The input slug tuned inductor was a commercial unit out of an old equipment. It has nine turns with a tap at five turns from ground and has a value of 4.7 uH as measured. A four turn link near ground end is wired as secondary. The tail transistor T3 is wired as a colpitts oscillator and an external BFO injection is thus not required. This added some simplicity to the final product detector shown below:

As you can see, Q4 has been removed as the product detector along with AF amplifier, an LM386N, provides about 30db gain which is more than enough. I used the LM386N in its low gain configuration i.e. I did not use a capacitor between its pin 1 and pin 8. As per my experience this basic configuration is really stable for this little device. It becomes unstable and noisy when we try to strain it by juicing up more gain out of it.

This completes our AF and IF stages. In the next post we will have a look at front end design of our receiver.

A DC Receiver Using Home Brewed Mixer IC

I am always fascinated with the sound of a direct conversion receiver. That sibilant, pure and dynamic sonic impact is just mesmerizing. I remember my first home brew receiver from "Solid state design" by Wes Haywards, using a CA3028A as mixer. It was almost three decades ago, in 1982 and still in use. Last weekend I decided to build another version of it using my home brewed mixer IC (as described in one of my previous post).

RECEIVER: The schematic diagram of complete circuit of the receiver is presented below. Input signal from antenna through a tuned circuit is fed to the mixer. I used an air core inductor with an ex-BC receiver tuning capacitor for the tuned circuit. However in case of nonavailability of the tuning capacitor, a varicap diode can be used as a replacement. The parts in the dotted square represents the home brewed mixer IC, I already have talked about. The mixer is simple, provides reasonable gain and requires little V.F.O. drive, which seems to be the primary requiments for such a portable QRP design. The output AF signal is filtered by a simple RC filter, which also acts as the only selectivity components in this design and the resultant signal is thus amplified by an amplifire built around Q4. This stage provides around 42db of audio gain. The amplifier can directly drive a high impedance headphones or a crystal earpiece. The over all sensitivity is very respectable and receiver can easily discern signals below 1uV.

However if you require to employ a speaker a small amplifier wired around LM 386N or a similar circuit can be used. A suitable circuit for such implementation is given below.

V.F.O.: The schematic diagram of the V.F.O. circuit is given in the following diagram. A single stage Hartley configuration employing a common BJT is used and it worked as expected, without any problems.

However I would like to state few points for V.F.O. stability:

1. Always build a V.F.O. in a separate shielded enclosure. For easy home construction you can make such enclosure from scrap double sided PCB laminates.

2. Always use a feed through capacitor and a series RF choke of about 200uH in series with the positive supply rail of the V.F.O. supply.

3. Anneal the V.F.O. inductor initially in boiling water for about 5 to 8 minutes and then allow its proper cooling before its use in the circuit. This eradicates the initial effect of mechanical stress on mettalic wire during wiring and let them set properly.

4. After building and initial testing seal the circuit with Araldite or similar adhesive to protect it from effects of humidity, temperature and vibrations.

5. Always make your capacitors in tuning and feedback circuit, to be a parallel combination of two or more units. This evenly distributes the RF current through them and reduces their heating and resultant thermal drift.

6. Always use black dot NP0 (C0G) or polestrene (styroflex) capacitors in your. V.F.Os.

I am sure that if you follow this basic philosophy many of the basic V.F.O. problems can be eliminated.

An IF Strip Using Homebrew ICs.

In a recent post I described a way to homebrew your own mixer ICs. It actally contain a differential single ended mixer which can be used in many ways as it has been stated in the end of that post. So, why not build a simple cascode amplifier with these sub circuits? It seemed to be a good idea and I tried it. A two stage IF stage amplifier with AGC was wired as shown in schematic diagram and was tested. The parts in dotted squares are homebrewed subcircuits. Using this two modules, I tried to prototype 4MHz IF amplifier strip of a two-stage cascade IF amplifier. Output tuning circuit of each stage is consists of an RFC of 4.7uH alongwith a capacitor of 330pF. These values can be altered for the frequency of interest or use. I used standard axial chokes for inductors but obviosly toroids are a better option. AGC is derived through a voltage doubler rectifier a portion of the output in 1N60 or 1N914 diodes,and a common PNP transistor.

The initial tests show a maximum IF gain of about 68dB, AGC  effectiveness started from an input of about 20dBuV EMF, that is quite an impressive figure for such a simple circuit with inexpensive devices. AGC versus gain characteristics are plotted in the following diagram: Indeed good designs can be evolved empirically with inexpensive components and simple thoughts.

SPOTTO:A High Performance DSB/CW Transceiver.

Over the past few months, attracted and inspired by the simplicity of direct conversion and DSB techniques I built several simple transceivers like that of Wee Willy and similar NE612+LM386 flavors including all the stuff I could Google on the net. Though technique of direct conversion appeals me for its added incentives of high performance versus simplicity and of its being less subjective to spurious responses and those designs don't entail the complexities of those high end superhets. But most of the designs including those mentioned earlier are of that Neolithic NE 602/LM 386 variants, more popular for their bad reputation and non-convincing performance. I, myself eschew the use of LM386 in serious receiver deign as much better and quieter alternatives are now available for cheap. Moreover I feel that for a given performance level, the fewer the components you use; the more critical to design each components becomes!

When I incepted the design of SPOTTO the focus was mainly on a simple and quieter high performance receiver with excellent sensitivity and dynamic performance combined with a simple DSB transmitter offering cleanest and moderate output for QRP use. The whole project is a collection of bench-marked building blocks put together to form a universal all band transceiver with the inclusion of a VFO/Synthesizer of user choice. The receiver uses a high level diode mixer which both is cheap and robust performing. It is usually imperative to properly terminate all ports of such mixer. Consequently I tried to do that in the simplest procedure (KISS way...!). Though it is not acceptable by the purists; but it certainly works for the QRPer. The mixer is followed by a W7EL low noise af amplifier with 50 ohms input impedance. A TL072 is chosen for the receiver audio for its low noise and small foot print, the prime essentiality for any serious and compact QRP design. A quieter TDA 7052 performs the final af amplification. The entire receiver is very lively with plenty of gain to give room filling audio. Further the TX/RX switching is kept to be minimum and simple. A conventional RF linear amplifier gives a reasonable 2.5W of RF output using a common BD139 on 40 meters but for higher bands a better device like 2SC1952 can be a much better choice. An additional module for CW enthusiasts is also available. No values for output RF filters have been given as there are G3RJV filters with values for all bands; already available on the club site. For those who expect multi-band operation, such filters can be constructed as three pin plug-in modules for easy band shifting. Many good DX were possible with the great sounding receiver during initial few evenings of use and several good compliments for its clean sounding signals were reciprocated. The transmitter is capable to produce Hall Effect that some amateurs seem to love a lot. I have assembled several copies of the project using VK3XU patchy board construction technique. It is easy to give it a go that way or you can ask editor, if the PCB for the project is available with the club store. PCB templates are given for those who wish to build their own. No PCB template is available for LPF and CW modules. They can be made on pigmy Vero board modules. LPF modules are made as three pin plug-in type modules for easy band changing. The use of suggested line filter circuit is strongly recommended to avoid common mode hum issues in case the rig is powered using a wall wart etc.

      

               

Home Brew Your Own Mixer ICs.

An integrated circuit (IC), is a semiconductor wafer on which thousands or millions of tiny resistors, capacitors, and transistors are fabricated. An IC can function as an amplifier, oscillator, timer, counter, computer memory, or microprocessor. A particular IC is categorized as either linear (analog) or digital, depending on its intended application.

Linear ICs have continuously variable output (theoretically capable of attaining an infinite number of states) that depends on the input signal level. As the term implies, the output signal level is a linear function of the input signal level. Analog or linear ICs invaded amateur designs in early seventies and were favored for cost and compatibility. There were numerous designs using them as mixers, IF amplifiers and AF amplifiers among others to say. Mixer ICs became popular for their ease of implementation in compact designs with predictable performance. The most popular devices being  NE 602/612, MC1496, SL640, CA3053, CA3046 and legendary CA3028.

Most of these devises are obsolete and out of production. The others are either hard to find and/or quite expensive; especially NE602/612 and MC1496. You have to shell a considerable wad of bucks to get one. Most western designs are designed around them, but I personally eschew the use of NE602/612 for its low IMD range leading to non-serious design contemplations. Lately I saw some designs using MC3362, TBA120, LA1185 and TA7358 as mixers. These devices are primarily developed for broadcast band radios and TVs, and are equally difficult to find in local markets. Thus; the non-availability of these items led me to look out for cheap alternative…….that is to fabricate my own integrated mixer ICs….!! Sounds impossible; but here is how I make them. 

Fabricating Your Own Mixer ICs: I have used almost all mixer ICs; afore-mentioned in my designs over the past three decades of home brewing. They produce almost the similar results in simple receivers. The one I love to play with is legendary CA3028, a single ended differential mixer. This IC is the most versatile device with a protean set of possibilities to go with. It can be used as a mixer, differential amplifier, a cascade amplifier with AGC, a balanced modulator, product detector or even a frequency multiplier. You can Google to download a detailed data sheet of this device. Even ever popular “Solid State Design” by Wes Haywards And Doug De Maw illustrates many designs around this popular device; which unfortunately is now out of production. The following diagram displays the internal diagram of IC CA3028A.

                   

All you have to do is take a good 8 Pin DIP socket and wire the entire circuit on it using discrete transistors and 1/8 watt resistors. The entire circuit can be wired as the above diagram using the component values mentioned. However for the transistors you can substitute BC547/548, 2N2222, 2N3904etc. for HF work as all of them perform very well up to about 50MHz. Beyond that any transistor with higher gain bandwidth product as BFR90 are quite suitable. The Q1 and Q2 should be matched units. You can use any of the following jigs to match transistors. All resistors are of 1% metal film type for best matching. 

                        

                                               

Give a resting period of few seconds for both transistors to be matched and do not touch them. The touch of finger can alter alter the junction temperature of the transistors and thus causing confusing results. The final home brewed mixer on the DIP socket this way can be used in any design as a standard IC circuit. It does not cost an arm and a leg and is far cheaper and easier alternative.

This home brew IC can be used as a mixer, a balanced modulator, product detector, IF amplifier or even as a frequecy multiplier as described in these suggestive drawings:

      A Balanced Mixer For DSB Generation.

                      An IF Amplifier with AGC.

 A Cascade IF Amplifier Stage of A Superhet.

               A Receiver Front End Mixer.

             An IF Amplifier With X-Tal Filter.

                                           Results of Two Tone Test of Home Made Mixer.

Idea from: http://www001.upp.so-net.ne.jp/jg1ead/ca3028modoki/ca3028modoki.html

                                            

The Power-Lore and QRP Philosphy.

QRP operation refers to transmitting at reduced power while attempting to maximize one's effective range. The term QRP derives from the standard Q-code used in radio communications, where "QRP" and "QRP?" are used to request, "Reduce power", and ask "Should I reduce power?" respectively. In practice it is a large and growing movement within the field of radio communication; both amateur and professional. The QRP fraternity has been growing exponentially and more and more QRP clubs are thriving world wide and in a scant two decades QRP operation has become a way of life for a plural wing of radio enthusiasts and is proliferating rapidly. It is evident now that working with QRP power levels isn’t a handicap; as it was once thought..!! It's just an arbitrary restriction of the one technical aspect of radio that has consistently worked against the interests of amateur operations. Set aside power, and you are left with skill, inventiveness, ingenuity, challenge, and enthusiasm that are very similar to the attractions of low power operation with added fun at lower cost along with the incentives of simplicity of designs and portability.

The Power Lore: Most amateurs use approximately 100 watts on HF and 50 watts on VHF/UHF. Though in some parts of the world, like the U.S. and Russia; they can use up to 1,500 watts. So many radio operators believe that higher the power the longer the distance you can work with. But it is a myth and nothing more than a philosophy of quasi lore. First; a little primer on power versus gain. Keep in mind that one S-unit on a properly calibrated receiver is equal to 6 db. To get a one S-unit gain you have to quadruple your power output. In other words, you would have to go to 20 watts to increase your signal strength by 6 db or one S-unit from your 5 watts. Conversely, to reduce your signal strength by 6 db or one S-unit, you need to go down to 1.5 watts from your 5 watts. Although it varies considerably due to many factors, one S-unit is about the minimum change in signal strength to be just noticeable. Here's a table comparing various power levels to 5 watts. An explanation follows.

Forget about logarithms and focus on the business end of the equation, the received signal. Signal strength is measured in S-points, which you can usually read directly from a meter on your radio. Your concern when transmitting is how many S-points you are generating at the receiving station. The more the better you believe. It may sound astounding but it is wrong scientifically. In the first place, if your signal is perfectly copiable at S-7, increasing the strength to S-9 achieves absolutely nothing other than hissing out atmospheric noise and grunting IMDs. If you want to increase your signal level by one S-unit, for example, look in the last column for 1.00 and you'll see you have to raise your power to 20 watts to get that change. I think it is very telling to look at the figures below 5 watts. Some folks think it is much 'greater' to get a QSO at 2.5 watts than with 5 watts. In reality there is only about 1/2 S-unit difference between the two powers, hardly noticeable at the receiving end. To drop your signal 2 full S-units requires going down to a little above 1/4 watt. Curious about the S-unit difference between say 100 watts and 1 watt? Just add the absolute values in the last column for 100 and 1 watts (2.17 + 1.17 = 3.34 S-units). I think the table helps explain a lot about why QRP can be so successful. Oh, although it is not in the table, the difference between 1,000 and 5 watts is 3.84 S-units. If a kW signal is S9, your QRP will be around S5 all other things being equal. You can have fun with the table and learn more about power and signal strength ratios. If a kW signal is S9, your QRP will be around S5 all other things being equal; and you are reasonably workable with quite visceral signals..!!

PWR	DB	S
100	13.01	2.17
50	10.00	1.67
40	9.03	1.51
30	7.78	1.30
20	6.02	1.00
10	3.01	0.50
5	0.00	0.00
4	-0.97	-0.16
3	-2.22	-0.37
2.5	-3.01	-0.50
2	-3.98	-0.66
1.25	-6.02	-1.00
1	-6.99	-1.17
0.5	-10.00	-1.67
0.25	-13.01	-2.17
0.125	-16.02	-2.67
0.0625	-19.03	-3.17

Don't believe….? Ok, let's look at the power ratio in action. Say you are transmitting with 5 watts and a station gives you a report of S-5. Now double your power to 10W and what happens? Your power output has increased by 3dB and the received signal has increased by the same 3dB, which is... wait for it.... one half of one S point. Double your power again, to 20W, and the received signal is now one whole S-point stronger. Double it again, to 40W and we are at 1.5 S points. Again, to 80W and we are at 2 S points improvement on our original 5 W signal. 80W is near enough to what your typical "100W" transmitter puts out, and by now you should see what little difference an additional 20W would make. In summary by going from 5W to 80W we have increased the received signal strength by all of two S points. The reverse is true; if you are copying an 80W station at S9 and he reduces power to 5W, you will still be copying him at S7.

But let's not leave it there. Start at 100W and add 3 dB at a time by doubling power- you go to 200, 400, 800, 1.6Kw. We doubled power 4 times, picking up 12 dB or.... wait for it.... 2 S points. Talk about diminishing returns! The only caveat is that the S-meters on most radios, if they are calibrated at all, are set for the standard S9 at 50uV input- at any other input, larger or smaller, they are notoriously inaccurate.

Thus the effectiveness of QRP communication, and the quality of QRP equipment, can be explained very easily with a little math. I hear you groaning, but it is very simple math and in fact you can witness its effectiveness, yourself in practice. A major factor in the continued growth and success of QRP is the cohesiveness of the QRP community. It is a community in all senses of the word, from local clubs to national organizations.

Less Frustation More Satisfaction: The downside of wanting to be a DXer if you only have an average station - which is all most of us can afford - is that the joy of making an unexpected DX contact is experienced less often than the frustration of when you don't work it. You only have to listen to the behavior of people in pile-ups to see what I mean. Many of them sound stressed. They don't sound as if they are having a good time. Is there pleasure to be had in shouting the last two letters of your call into a microphone for half an hour, especially if at the end of it you have nothing to show for it? I don't think so. A few of the high-power guys like to claim that it's their ears that do all the work during a QRP contact. "QRPers brag about working DX with milliwatts but it's the guy other end who does all the work to make the contact" is a typical comment. But if you believe that QRP operators brag about making a contact, you don't understand the QRP mentality at all. What the QRP op feels is a sense of wonder that such a tiny amount of power can propagate a signal to the other side of the world. The QRO op at the other end can feel that too. And from the comments I sometimes get ("your QRP is doing great" etc.) I think that the grumpy guys who think QRP just makes work for them are in the minority.

Further Thoughts: The reason I think all this is important is that QRP amateur radio is something which is achievable by all. A high power, DX-capable station is achievable by a minority. Some simply can't afford it. Others don't have the opportunity to erect big antennas, or simply don't want to displease family members or annoy the neighbors. Be honest: how many non-hams think antennas are attractive? QRP is cleaner than QRO and does not advocate RF pollution. Advancements in RF practice are on and new avenues for QRP and QRPp (Very low power communications) are opening. Several weak signal digital modes like CCW, JAT63/65, WSPR,WSJT and QRSS are becoming popular among designers and operators. The program WSJT, is created by Nobel Prize winning Princeton professor, Joe Taylor, K1JT. Its purpose is to send and receive various weak signal modes for meteor and ionospheric scatter as well as EME (moonbounce) on VHF and UHF bands. And, it’s both free and well supported. On the other hand QRSS is a derivative of the CW Q-Signal QRS for "Please lower your code speed". By using extremely slow CW, it is possible to use a computer sound card and special software to extract CW characters from below the audible noise floor. Morse code element lengths of 10 to 30 seconds (or even longer) per dot are commonly used.

Amateur VLF operators have used QRSS techniques to span the Atlantic at 136 KHz and to receive very weak VLF beacon transmissions from distant locations. By adopting these same techniques, QRP operators can push the envelope of very low power HF communications.

                    A 6 milliwatt QRSS contact in action.